The present invention relates to signal synchronization in a communication system and, more particularly, to a device for and a method of signal synchronization in an orthogonal frequency division multiplexing (OFDM) system using pseudo-random noise (PN) guard intervals (GIs).
In a communication system, a base station may process (e.g., encode and symbol-map) data to obtain modulation symbols, and may further process the modulation symbols to generate a modulated signal. The base station then transmits the modulated signal via a communication channel. The communication system may use a transmission scheme whereby data are transmitted in frames, with each frame having a particular time duration. Different types of data, such as traffic/packet data, overhead/control data and pilot, may be sent in different parts of each frame.
A terminal in the system may not know which base stations, if any, near its vicinity are transmitting. Furthermore, the terminal may not know the start of each frame for a given base station, the time at which each frame is transmitted by the base station, or the propagation delay introduced by the communication channel. The terminal may perform signal acquisition to detect for transmission from base stations in the system and to synchronize to the timing and carrier frequency offset of each detected base stations of interest. By performing the signal acquisition process, the terminal can ascertain the timing of each detected base station and can properly perform the complementary demodulation for that base station.
Due to non-ideal channel effects, for example, multi-path reflection and multi-path fading, signals transmitted from base stations to terminals via physical channels (such as air) in a communication system may be distorted. An Orthogonal Frequency Division Multiplexing (OFDM) system that uses multi-carrier modulation technique may effectively solve the problems caused by multi-path reflection effect. In an OFDM system, a simple first-order equalizer at a terminal may equalize constructive and destructive interferences from multi-path reflection effect. FIG. 1A is a schematic diagram of an exemplary symbol format in a conventional OFDM system. Referring to FIG. 1A, a terminal (not shown) may cyclically copy a Guard Interval (GI, which has a length of Tg) from a useful OFDM symbol (which has a length of Tu with a number of “N” samples) and then combine the copied GI with the useful OFDM symbol to form a complete symbol “m”. Inter-Symbol Interference (ISI) introduced by multi-path at terminals in the OFDM system may be avoided while the maximum delay τmax of a channel is smaller than Tg. The terminals then may remove GI and extract the useful OFDM symbol, perform Fast Fourier Transform (FFT) on the useful OFDM symbol and estimate a channel frequency response in accordance with pilot carriers. A first-order equalizer may subsequently compensate channel effects to estimate the transmitted data.
A Time Domain Synchronization (TDS) OFDM system may perform fast synchronization and reduce system resources (e.g., bandwidth). FIG. 1B is a schematic diagram of an exemplary symbol format in a TDS-OFDM system. Referring to FIG. 1B, in a TDS-OFDM system, a representative symbol “m” may include a GI with a Pseudo-random Noise (PN) sequence, which may exhibit desirable autocorrelation characteristics. A sliding correlator for the received signal and the local PN sequence may be used to rapidly perform symbol timing synchronization and estimate a channel impulse response. An output of the sliding correlator may include the channel impulse response, which indicates the strength and the location for each of the paths in the multi-path channel. The output of the sliding correlator may also include the information on symbol timing. For example, the location of the peak value of the sliding correlator output may represent the coarse symbol timing (CST). Moreover, channel impulse response for the multi-path channel can be estimated in accordance with the output of the sliding correlator. In other words, scattered pilot carriers are no more required in the TDS-OFDM system. Consequently, bandwidth consumption of the total system may be significantly reduced.
However, the desirable autocorrelation characteristics of a PN sequence may be susceptible to carrier frequency offset (CFO), which may attenuate the peak value of an output of a sliding correlator. FIGS. 2A and 2B are schematic diagrams illustrating phase distribution in the autocorrelation of a PN sequence due to a relatively small CFO and a relatively large CFO, respectively. The peak value of a sliding correlator may equal an average sum of a number of “M” linear phases each being proportional to the CFO, as shown in FIGS. 2A and 2B. Referring to FIGS. 2A and 2B, the CFO may affect the magnitude of the autocorrelation of a sliding correlator, which in turn may attenuate an output peak value of the sliding correlator. When the CFO reaches an integer-fold of an integral CFO (ICFO) increment, the peak value of an output of the sliding correlator may drop to zero. The ICFO increment may be defined as 1/(MTs), where “M” is the number of samples in a PN sequence and Ts is the sample length. Consequently, the larger the carrier frequency offset, the more significant attenuation of the output peak value of the sliding correlator.
FIG. 2C is a diagram illustrating peak values of a sliding correlator at various CFOs. The horizontal axis of FIG. 2C represents CFOs in carrier spacing (fsub), and the vertical axis represents the magnitude of the autocorrelation outputs of the sliding correlator in the square of millivolt ((mV)2). Referring to FIG. 2C, given a sub-carrier spacing (fsub) of two (2) kilohertz (KHz), N=3780 samples and M=255 samples, for initial CFOs ranging from −100 to 100 carrier spacings (fsub), i.e., −200 to 200 KHz, the ICFO may be calculated below.ICFO increment=1/(MTs)=(N/M)fsub, with fsub=1/(NTs)That is, ICFO increment=3780/255(fsub)≅14.82(fsub)
As illustrated in FIG. 2C, the peak values of the sliding correlator may equal approximately zero at CFOs which are integer-folds of 14.82 carrier spacings (fsub). In an OFDM system, the location of a maximum output peak of the sliding correlator for the received signal and the local PN sequence detected in an OFDM symbol period may correspond to a coarse symbol timing (CST). If the value of the maximum output peak of a sliding correlator is attenuated, information regarding CST may be lost.
Sliding correlation with differential demodulation may be used to prevent the autocorrelation characteristics of a PN sequence from being destroyed by CFOs. FIG. 3 is a block diagram of a conventional communication system 1 using sliding correlation with differential demodulation. Referring to FIG. 3, the communication system 1 may include a differential modulator 12 in a base station and a synchronization unit 14 in a terminal. The synchronization unit 14 may include a differential demodulator 141 and a sliding correlation & peak detection device 142. The synchronization unit 14, which is insensitive to initial CFO, may differentially demodulate a received signal in the differential demodulator 141 and then perform sliding correlation and peak detection in the sliding correlation & peak detection device 142 for CST. The communication system 1 may alleviate the CFO issue, however, the system structure may be complicated because the differential modulator 12 is required for each base station in the system 1. Accordingly, in communication systems that employ other modulation techniques than the differential modulation, a synchronization unit like the unit 14 at a terminal side cannot differentially demodulate a received signal and provide desirable sliding correlation function.
It may therefore desirable to have a device for and a method of signal synchronization in a PN GI-based OFDM system, which may provide sliding correlation that is relatively robust to CFO and may be implemented in a relatively simple structure.